1. Field of the Invention
The present invention relates to a voltage to current converter suited for an active filtering circuit.
2. Description of Prior Art
To construct an active filter on an integrated circuit, a voltage to current converter is an important element. FIG. 10 shows an example of an active filter formed on an integrated circuit.
It is difficult to achieve a condenser of a large capacity onto an integrated circuit. Practically, several ten pF is adequate and several thousand pF is the limit even if a portion occupied on a chip area is considerably allowed. For this reason, it is impossible to realize a filter with a low cut-off frequency by increasing the capacity of the condenser. Also, resistances formed on the integrated circuit are diffusion resistances whose values have poor absolute accuracy and large temperature dependence, although the ratio of the resistance values between respective resistances is constant.
Therefore, when an active filter is formed on an integrated circuit, such restriction must be considered fully. An active filter shown in FIG. 10 is made up of in consideration of the above restriction to provide a sufficient accuracy even if the cut-off frequency is low.
In FIG. 10, reference numerals 101 and 102 show PNP type transistors, the emitter of the transistor 101 is connected to one end of a resistance 103, and the emitter of the transistor 102 is connected to one end of a resistance 104. The other end of the resistance 103 and that of the resistance 104 are coupled, and its junction is connected to the collector of a PNP type transistor 105 serving as a current source. A V-I (voltage to current) converting circuit 151 is constructed by the transistors 101, 102 and the resistances 103, 104.
The emitter of a transistor 105 is connected to a power supply terminal 106 of power supply voltage Vcc. The base of the transistor 105 and that of a PNP transistor 107 are commonly connected, and the base of the transistor 107 is connected to the collector of the transistor 107 to form a current-mirror circuit. The emitter of the transistor 107 is connected to the power supply terminal 106, and the collector of the transistor 107 is connected to that of an NPN type transistor 108.
The base of the transistor 108 and that of an NPN type transistor 110 are connected in common, and the base of the transistor 110 is connected to its collector to form a current-mirror circuit. The emitter of the transistor 108 is connected to an earth terminal 109. The emitter of the transistor 110 is connected to the earth terminal 109. The collector of the transistor 110 is connected to one end of a resistance 111. A reference voltage source 112 is connected between the other end of the resistance 111 and the earth terminal 109.
By the reference voltage source 112, a current 2 : flows into the resistance 111 and the transistor 110. Because the transistors 110 and 108 are current-mirror coupled, the current 2I.sub.10 equal to that flowing into the transistor 110 flows into the transistor 108. Since the transistors 108 and 107 are connected in series, a current equal to that flowing into the transistor 108 flows into the transistor 107, and the current 2I.sub.10 current-mirror coupled to the transistor 107 flows into the transistor 105. As a result, the transistor 105 operates as a constant current source having the current value 2I.sub.10. The base of the transistor 101 is connected to an input terminal 113. The base of the transistor 102 is connected to an output terminal 133. The collector of the transistor 101 is connected to the anode of a diode 114 and to the base of an NPN type transistor 121. The collector of the transistor 102 is connected to the anode of a diode 115 and to the base of an NPN type transistor 122. The cathodes of the diodes 114 and 115 are coupled to the anode of a diode 116. The cathode of the diode 116 is coupled to the earth terminal 109.
A multiplier 152 is composed of the diodes 114, 115 and the transistors 121, 122.
The emitters of the transistors 121 and 122 are commonly coupled, and its junction is connected to the collector of a transistor 123 acting as a current source. The emitter of the transistor 123 is connected to the earth terminal 109. The base of the transistor 123 is commonly coupled to the base of a transistor 124 and that of a transistor 125, and the base of the transistor 125 is coupled to its collector to form a current-mirror circuit. A terminal 130 is led from the collector of the transistor 125. On the other hand, a terminal 131 is derived from the junction of the reference voltage source 112 and the resistance 111. A resistance 132 is externally provided between the terminals 130 and 131.
By the reference voltage source 112, a current I.sub.11 flows into the externally provided resistance 132 and the transistor 125. Since the transistors 125 and 123 are current-mirror coupled, the constant current 2I.sub.11 flows into the transistor 123.
The collector of the transistor 121 is connected to that of a PNP type transistor 126 and to the base of an NPN type transistor 128. The emitter of the transistor 126 is coupled to the power supply terminal 106. The collector of the transistor 122 is connected to the collector of a PNP type transistor 127. The base of the transistor 127 is commonly connected to that of the transistor 126, and the base of the transistor 127 is connected to its collector to form a current-mirror circuit. The emitter of the transistor 127 is connected to the power supply terminal 106.
A condenser 129 is connected between the junction of the collector 121 and the base of the transistor 128 and the earth terminal 109. The collector of the transistor 128 is connected to the power supply terminal 106. The emitter of the transistor 128 is connected to the collector of the transistor 124, and an output terminal 133 is led from the emitter of the transistor 128. The emitter of the transistor 124 is connected to the earth terminal 109. The bases of the transistors 124 and 125 are commonly connected to form a current-mirror circuit. The transistor 124 operates serving as a current source for the emitter-follower transistor 128.
An operation of the above-mentioned active filtering circuit will be described.
An input signal from the input terminal 113 is supplied to the base of the transistor 101 of the V-I converting circuit 151. The output signal of the emitter-follower transistor 128 is fed back to the base of the transistor 102. The differential output of the V-I converting circuit 151 is given to the multiplier 152. The output of the multiplier 152 is converted into a single-end output by the current-mirror circuit composed of the PNP type transistors 126 and 127 and given to the condenser 129. A signal obtained from terminal voltage of the condenser 129 is taken out from the output terminal 133 through the emitter-follower transistor 128.
Assuming that an input signal supplied to the input terminal 113 and an output signal produced from the output terminal 133 are v.sub.in and v.sub.out, respectively, the emitter voltage of the transistor 101 becomes (v.sub.in +V.sub.BE), and the emitter voltage of the transistor 102 takes (v.sub.out +V.sub.BE). As a result, assuming that the resistance values of the resistances 103 and 104 are R.sub.e, respectively, a current ((v.sub.in -v.sub.out)/(2.Re)) flows through the resistances 103 and 104. Output currents i.sub.11 and i.sub.12 of the V-I converting circuit 151 are obtained approximately by the following equations on the assumption that the current 2I.sub.10 flows into the transistor 105: EQU i.sub.11 =I.sub.10 -(v.sub.in -v.sub.out)/(2.R.sub.e) (51) EQU i.sub.12 =I.sub.10 +(v.sub.in -v.sub.out)/(2.R.sub.e) (52)
Assuming that the current I.sub.11 flows into the transistor 123, it is known that output currents i.sub.13 and i.sub.14 of the multiplier 152 become as follows: EQU i.sub.13 =I.sub.11 -(I.sub.11 /I.sub.10).(v.sub.in -v.sub.out)/(2.R.sub.e) (53) EQU i.sub.14 =I.sub.11 +(I.sub.11 /I.sub.10).(v.sub.in -v.sub.out)/(2.R.sub.e) (54)
Therefore, a charge current i.sub.c for the condenser 129 is given as: EQU i.sub.c =i.sub.14 -i.sub.13 =(I.sub.11 /I.sub.10).(v.sub.in -v.sub.out)/R.sub.e ( 55)
Now, assuming that (I.sub.11 /I.sub.10)/R.sub.e is a transfer conductance Gm, Equation (55) can be expressed by: EQU i.sub.c =Gm (v.sub.in -v.sub.out) (56)
Now, the input voltage v.sub.in, the output voltage v.sub.out and the current i.sub.c are instant values, respectively. However, to obtain the transfer function of this circuit by defining V.sub.in (s), V.sub.out (s) and I.sub.c (s), respectively, as the function of s(=j.omega.), EQU I.sub.c (s)=Gm(V.sub.in (s)-V.sub.out (s)) (57)
is established. Also, assuming that the electrostatic capacity of the condenser 129 is C.sub.o, EQU V.sub.out (s)=I.sub.c (s)/(s C.sub.o) (58)
is established.
Therefore, by obtaining V.sub.out (s) from Equations (57) and (58), ##EQU1## is obtained. From the above equation, it is indicated that this circuit exhibits a first-order low-pass filtering characteristic.
The cut-off frequency of this circuit is defined by the electrostatic capacity of the condenser 129 and the transfer conductance Gm. In an integrated circuit, a considerably accurate and less temperature-dependent capacity can be realized by the use of an insulating film such as a nitride film, for example. Consequently, if a correct transfer conductance can be realized, a correct filtering characteristic can be provided.
The transfer conductance Gm is defined as mentioned above by: EQU Gm=(I.sub.11 /I.sub.10)/R.sub.e ( 60)
Here, the current ratio (I.sub.11 /I.sub.10) is defined by the internal resistance 111 and the externally provided resistance 132 rather than a reference voltage source 112. Assuming that the resistance values of the resistances 111 and 132 are R.sub.111 and R.sub.132, respectively, since EQU I.sub.11 /I.sub.10 =R.sub.111 /R.sub.132 ( 61)
the transfer conductance Gm is given as: EQU Gm=R.sub.111.R.sub.e /R.sub.132 ( 62)
Now, since the resistance values R.sub.111 and R.sub.e are diffusion resistances formed in the integrated circuit, the absolute accuracy is inferior but the resistance ratio can be kept constant. Assuming that the resistance ratio of the resistances R.sub.111 and R.sub.e is N, EQU Gm=N/R.sub.132 ( 63)
is established.
As a result, the transfer conductance does not depend on the absolute accuracy of the resistances, thereby to provide an accurate transfer conductance as well as to achieve a filtering circuit with a correct cut-off frequency.
In this way, the above-mentioned active filtering circuit is capable of providing an accurate characteristic, since the cut-off frequency does not depend on the absolute values of the resistances on the integrated circuit and is defined by the relative ratio of the resistances on the integrated circuit and the externally provided resistance. In addition, by increasing the resistance value R.sub.132 of the resistance 132 and decreasing the resistance ratio N, a low cut-off frequency can be obtained without making the capacity of the condenser extremely large.
However, in the above-described active filtering circuit, offset voltage .DELTA.V.sub.o as equivalently shown at a voltage source 161 between the collector of the transistor 101 and the anode of the diode 114 in FIG. 10 due to mismatching in respective saturation currents of the diode pair consisting of the diodes 114, 115, the transistor pair consisting of the transistors 121, 122 and the transistor pair consisting of the transistors 126, 127; respective Allee's effects of the transistor pair consisting of the transistors 121, 122 and the transistors 126, 127 constituting the current-mirror circuit; and respective current amplification gains of the transistors 126, 127 forming the current-mirror circuit and the emitter follower transistor 128. By the offset voltage .DELTA.V.sub.o thus generated, offset voltage V.sub.off is generated at the output terminal 133. This offset voltage V.sub.off is represented approximately by the following equation: ##EQU2## where V.sub.T is thermal voltage kT/q (k: Boltzmann constant, T: absolute temperature, q: elementary electric charge) and has a value of 26 mV at a normal temperature. If .DELTA.V.sub.o &lt;&lt;V.sub.T is established, Equation (64) is given as: EQU V.sub.off .apprxeq.R.sub.e.I.sub.10..DELTA.V.sub.o /V.sub.T ( 65)
Now, on the assumption of R.sub.e =10 K.OMEGA., I.sub.10 =100 .mu.A and .DELTA.V.sub.o =3 mV, for instance, the offset voltage V.sub.off is about 120 mV. The maximum value of signal levels, which can be dealt with this circuit, is restricted by the restance value R.sub.e of the resistances 103 and 104 and the current value I.sub.10, and 4 R.sub.e.I.sub.10 is the maximum value of the signal levels to be handled in terms of peak to peak value. Therefore, in this case, it is 4 V.sub.p-p. In this manner, if the voltage .DELTA.V.sub.o is determined with respect to the maximum value of the signal levels to be handled in this circuit, the offset voltage V.sub.off is produced at a constant rate.
Such generation of the offset voltage V.sub.off does not pose a serious problem, since an input signal is supplied through a coupling condenser in a filtering circuit using for suppression of spurious components, for example. However, there is a case where such offset voltage V.sub.off becomes a great problem. An example in which the offset voltage V.sub.off brings about a big problem will be described specifically hereunder.
FIG. 11 shows an encoding circuit of an autio-noise reduction circuit for a compact cassette tape recorder. An audio signal recorded through the encoding circuit shown in FIG. 11 is reproduced into an original voice signal by a decoding circuit having a symmetrical characteristic with this circuit.
In FIG. 11, reference numeral 201 indicates an input terminal. An input signal from the input terminal 201 is supplied to a variable high-pass filter 202 and an adder 205. The output of the adder 205 is taken out from an output terminal 206. In the variable high-pass filter 202, its cut-off frequency varies by control voltage from a level detector 204 and the cut-off frequency increases with the increase of a signal level. The output of the variable high-pass filter 202 is supplied to the adder 205 and a high-frequency region weighting circuit 203. The output weighting circuit 203 is given to the level detector 204.
In the absence of a signal, the cut-off frequency of the variable high-pass filter 202 is in the lowest state. By the adder 205, the input signal and the signal passing the variable high-pass filter 202 are added so that the gains in the middle and high frequency regions are raised by approximately 10 dB. On the other hand, the middle and high frequency regions elevated at the time of recording is attenuated by about 10 dB in a decoding circuit on the reproduction side. As a result, noises in the middle and high frequency regions are attenuated by approximately 10 dB. With the increase of the signal level, the cut-off frequency of the variable high-pass filter 202 is raised and the frequency characteristic of the circuit approaches a flat state. Since the decoding circuit on the reproduction side approaches a flat state, a noise reduction effect decreases. However, in this state, no noise is sensed because a masking effect by the signal works.
The high-frequency region weighting circuit 203 is a filtering circuit and acts to increase the cut-off frequency of the variable high-pass filter 202 when an input frequency increases. This circuit prevents the frequency characteristic in the middle and high level regions from being elevated in the high frequency region to provide a flat characteristic or the one with a slightly lowered high-frequency region.
In the case where the above-mentioned enoding circuit for the audio-noise reduction circuit is realized in an integrated circuit together with the high-frequency region weighting circuit 203, the generation of the offset voltage V.sub.off will become a serious problem. Clearly, in FIG. 11, the level detector 204 must correctly detect signal levels extending over a considerably broad range. With the offset voltage V.sub.off generated, an error may occur in the control voltage given from the level detector 204. Although it is conceived that a coupling condenser is inserted between the high-frequency region weighting circuit 203 and the level detector 204, this may bring about the increase of pins led from a package, since an externally provided condenser becomes necessary.
An object of the invention is, therefore, to a voltage to current converter capable of constructing an active filtering circuit free from the generation of offset voltage and having an accurate direct current transmission characteristic.